It is the task of d.c. voltage converters to transform a direct voltage applied at their input into a direct voltage which is output at a different value, and to do so most efficiently. The output value may be greater or smaller than the input value and may be adjustable. In a d.c. voltage converter the direct voltage input, first, is transformed with the help of switching stages into alternating voltage having a rectangular waveshape. During the switch-on period “chopped” direct voltage is stored in the form of magnetic energy in a choke. During the switch-off period, it acts as a self-induction voltage at the output of the d.c. voltage converter. D.C. voltage converters operating according to this principle are referred to as choke converters. They have their inputs and outputs galvanically separated. It is known in the art to use transformers to achieve separation of potential. Here, the induced voltage occurs at the secondary winding, and the voltage transformation can be determined by the number of windings. FIG. 1 is a block diagram illustrating a transformer-type converter used as d.c. voltage converter. FIG. 1 depicts the basic elements of a transformer-type converter, including an input switching stage 10, a power transformer 12, a rectifier circuit 14, and an output filter 16. A distinction is made in the art between single phase d.c. converters and push-pull voltage transformers. A single phase d.c. converter may be regarded as being a simple electronically controlled switch, whereas switch-over operations occur with push-pull voltage transformers, and a transformer 12 having two primary windings may be required. Push-pull transformers can be derived from two single phase d.c. converters connected in parallel. The electronic switch-over is accomplished by two switching stages, and current always flows through one of the two primary windings. The invention relates to the field of push-pull voltage transformers.
In practice, such push-pull voltage transformers are employed in switch mode power supply, such as server architecture for telecommunications applications, in PCs, industrial applications, and many other situations. The invention is especially advantageous in distributed energy supply systems where several stages are connected in succession. In new server architectures, for instance, power supply units are used in which the mains voltage, to begin with, is converted into a bus voltage of some 48 to 50 V. A second conversion to +12 V, for example, then takes place within the server sub-system. The specific voltages required for the various components, such as the microprocessor, RAMs, etc. are produced locally by so-called voltage regulator modules which are connected to the 12 V rail.
Each of the converting stages must operate as efficiently as possible in view of the series connection of different power stages. For maximum efficiency to be obtained with switch mode power supply, optimization in terms of energy loss is required regarding each and every source thereof. Energy losses depend not only on the type of converter or transformer chosen, be it a single phase d.c. converter or a push-pull voltage transformer, but are determined decisively by the mode of operation of the rectifier circuit.
Making good use of the inductivities as well as driving the transducer in positive and negative directions present great advantages of push-pull voltage transformers. Another advantage is the great efficiency of the transformer and the high output power attained.
FIG. 2 is a schematic circuit diagram of a push-pull voltage transformer provided with Schottky diodes for rectification at the secondary side. The push-pull forward converter illustrated in FIG. 2 is known; its function is described, for example, in Billings, Keith “Switch Mode Power Supply Handbook”, 2nd edition, McGraw-Hill, New York, 1999. This converter comprises a power transformer 18 having a primary side 18a and a secondary side 18b. The primary side 18a and the secondary side 18b each comprise two winding sections. For driving purposes, two power transistors 20, 22 are associated with the two winding sections at the primary side 18a. Two secondary diodes 24 and 26 are associated, respectively, with the two winding sections at the secondary side 18b. These diodes are connected to an output filter stage made up of a storage choke 28 and a storage capacitor 30, as may be seen in FIG. 2. The power transistors 20, 22 are driven, for instance, by a control, IC (not shown).
As the transistor 22 is driven, current will flow through the associated winding section of the power transformer 18 and also through the transistor 22. The polarity of the associated winding section at the secondary side 18b of the power transistor 18 causes the diode 26 to cut off. At the same time, voltage is induced also in the other winding section at the secondary side 18b, thus causing current to flow through the diode 24 via the storage choke 28. When a sufficient amount of energy has been transmitted from the primary side to the secondary side, the transistor 22 blocks. During the next cycle the transistor 20 is driven. The current now flowing through the second winding section of the primary side 18a causes reversal of the polarity of the associated winding at the secondary side 18b. Diode 24 blocks, while diode 26 is conducting, thus permitting current to flow through the choke 28, as during the first cycle. To make sure that the two power transistors 20, 22 will not be conducting at the same time, a compulsory break, the so-called freewheeling phase, is provided between the first and second cycles described above. During this freewheeling phase, the electric circuit at the secondary side 18b of the transformer 18 is formed of the storage choke 28, the storage capacitor 30, the two conducting diodes 24 and 26, and the connected load (not shown in the drawing).
FIG. 3 presents idealized waveshapes of the output voltages u01, U02 at the two winding sections of the secondary side 18b of the power transformer 18, forward currents i01, i02 through the diodes 24 and 26, and the output current i0 through the storage choke 28.
A positive voltage u01 is generated during the first time interval from t1 to t2. Diode 24 is conducting. The overall output current i0 is passed through the same and through the upper secondary winding section of the power transformer 18 toward the output. The rise in output current is determined by the voltage difference u01−u02 (output voltage) and the sum of the inductivities of the secondary circuit.
The second time interval from t2 to t3 corresponds to the so-called freewheeling phase. The output voltages u01 and u02 of the transformer 18 are zero. The current i0 is determined by the inductivities of the secondary circuit. If the upper and lower winding sections at the secondary side 18b are identical the output current i0 is divided in two. Each of the diodes 24, 26 will carry one half of the output current i0. During this time interval the output current drops.
During the third time interval from t3 to t4 a positive voltage u02 is produced and diode 26 is conducting. The resulting behavior corresponds to that of the first time interval.
During the last time interval from t4 to t5 of period T both power transistors 20, 22 are turned off. The voltages u01 and u02 once again are zero, which corresponds to the freewheeling phase.
In the embodiment of the d.c. voltage converter shown in FIG. 2 the secondary rectifier is embodied by diodes. The rectifier diodes produce losses which depend on the forward voltage of the diodes 24, 26 and are composed of forward losses and switching losses of the diodes.
The forward loss PDC of a diode is given by the product of its forward voltage drop uF and its forward current iD (see also FIG. 4)PDC=uF·iD.
The forward voltage rises as the load increases; it lies between 0.5 V and 1.5 V, depending on the type of diode provided. If the transducer output voltage is 3.3 V, for example, which would correspond to a processor voltage, as much as 30% of the voltage will drop at the rectifier diodes. With higher transducer output voltages, e.g. 48 V in telecommunications applications, the voltage drop at the diodes is comparatively less, but still not negligible.
The switching loss of a diode can be estimated by the following equation:PDS=QF·û·f
where QF is the recovered load during the fall time of the reverse current of the diode, f is the reciprocal value of period T, and-is the peak value of the diode turnover voltage.
A reduction of the forward loss discussed above can be achieved only by reducing the voltage drop.
One solution resides in the use of a MOSFET connected in parallel with the diode. That is shown in FIG. 4. The MOSFET is turned on when current in forward direction is applied to the diode, and it is turned off when the current is reversed. This is called synchronous rectification. The diode, such as diodes 24, 26, in a circuit may be replaced by a MOSFET. If the MOSFET used is of vertical structure its antiparallel diode or inverse diode (body diode) is utilized. This is illustrated in FIG. 5 which shows like members identified by like reference numerals as in FIG. 1. FIG. 5 diagrammatically illustrates the replacement of the diode-type rectifier 14 by a synchronous rectifier circuit 32 on the basis of MOSFETs.
With reference to FIGS. 4 and 5 it becomes clear that the voltage drop uDS at the MOSFET is determined by the switch-on resistance RDS(ON) of the MOSFET and the actual drain current which must equal the diode current iD. The following must apply if the energy loss is to be reduced:|uDS|=|RDS(ON)·iD|>uF.
The forward loss, therefore, can be reduced by selecting a MOSFET of which the forward resistance RDS(ON) is small.
A control signal is required for the MOSFET to be switched on and off. Generating the control signal has a decisive influence on the switching behavior. Moreover, the energy losses in this circuit must be taken into consideration. There are various known methods of driving synchronous rectifiers comprising a MOSFET, and they may be roughly classified as self-controlled, IC controlled, and current controlled.
FIG. 6 is a simplified diagram of the secondary side of a self-controlled synchronous rectifier circuit. Again the same reference numerals are used to designate members corresponding to those of FIG. 2. The power diodes 24, 26 shown in FIG. 2 are replaced by MOSFETs 34 and 36, respectively. The first winding section at the secondary side 18b is designated LS1, and the second winding section is designated LS2.
In the case of the self-controlled synchronous rectifier according to FIG. 5 the output voltage of the power transformer 18 is used to control the MOSFETs 34, 36. This circuit has the advantage of necessitating only little expenditure in circuitry since no additional driver circuits are needed to drive the MOSFETs 34, 36.
With reference to FIG. 3, the output voltage u01 at the first winding section LS1 is positive during the time interval from t1 to t2, while u02 is negative at the second winding section LS2. With these conditions, the p-channel MOSFET 34 is switched on because its gate voltage is negative. Corresponding switching behavior is true of the p-channel MOSFET 36 during the time interval from t3 to t4; MOSFET 34 is turned off, while MOSFET 36 is turned on. Operation of the MOSFET switches 34, 36 during these two time intervals is satisfactory. During the freewheeling phase, however, no control voltage is generated. The current flows through the inverse diodes of the MOSFETs, whereby higher forward losses are produced than necessary.
Furthermore, it is disadvantageous with the self-controlled synchronous rectifier design that p-channel MOSFETs are needed which are much more expensive and have a higher forward resistance than comparable n-channel MOSFETs. Moreover, the transformer output voltage range is restricted due to the gate voltage of the MOSFETs 34, 36. It must be higher than the threshold voltage and lower than the maximum permissible gate voltage of the MOSFETs of approximately 30 V.
FIG. 7 is a circuit diagram of the secondary side of a synchronous rectifier with IC drive, corresponding members being designated by the same reference numerals as in FIG. 6. With this synchronous rectifier design, driver ICs 38, 40 are provided to drive the MOSFETs 34, 36. There are only a few manufacturers who offer such specific driver ICs for synchronous rectifiers. The IC components 38, 40 scan the secondary voltages of the transformer 18, and the MOSFETs 34, 36 are turned on or off, depending on the potential profile. The electronic control assures synchronous switching on and off of the synchronous rectifier. However, the scarce commercial availability and the relatively high costs and greater expenditure involved in connecting and feeding the driver ICs 38, 40 are points against making use of driver ICs 38, 40.
FIG. 8, finally, is a circuit diagram of the secondary side of a current controlled synchronous rectifier, showing only the upper part of the secondary side 18b which includes the first secondary winding section LS1. The structure of the lower part, including the second secondary winding section LS2 is mirror inverted.
In the current controlled synchronous rectifier, the power MOSFET 34 (and 36, too, not shown in FIG. 8) is controlled through a current transformer 42. The current transformer 42 is connected in series between the upper winding section LS1, at the secondary side 18b of the power transformer 18 and the MOSFET 34 and comprises a primary winding 42a and a secondary winding 42b. The secondary winding 42b is connected to the gate of the MOSFET 34 by way of a voltage divider composed of two resistors 44, 46.
When the transformer 18 is controlled such that current flows through the winding section LS1, of the secondary side 18b current also will flow through the inverse side of the MOSFET 34, the current transformer 42 thus generating current in its secondary winding 42b. This current brings about a voltage drop across the resistor 46 of a magnitude equal to the gate voltage of the MOSFET 34. The value of the voltage drop is adjustable by the ratio between the two resistors 44, 46.
MOSFET 34 is turned on during the time interval from t1, to t2, i.e. it is on also during the freewheeling phase. When current flows in the opposite direction the output voltage at the secondary side 42 of the current transformer 42 becomes negative and MOSFET 34 turns off. The second section LS2 of the power transformer 18 and the second MOSFET 36 (not shown in FIG. 8) behave accordingly, yet with opposite sign.
Controlling synchronous rectifiers by means of current transformers, such as illustrated in FIG. 8, has certain disadvantages. On the one hand, a MOSFET requires a great current pulse to be turned on, which means that the windings ratio n2/n1 of the current transformer must be low. In the switched-on state, on the other hand, the gate current of the MOSFET is negligible, which means that a high windings ratio n2/n1 of the current transformer is required.
A description of prior art similar to what has been described above, but relating to a single phase forward transformer with synchronous rectification, including a current transformer, is to be found in “Synchronous Rectification Circuit Using A Current Transformer” by Y. Kubota et al., NTELC Conference Proceedings, September 2000, pages 267 to 273.
It is an object of the invention, starting from the state of the art as described above, to indicate a synchronous rectifier circuit for a push-pull voltage transformer that attains the fastest possible switching of the metal oxide semiconductor field effect transistors (MOSFETs) while, at the same time, causing the least possible power dissipation. This aim is to be reached, above all, by generating a higher switch-on current for the MOSFETs so as to keep the time of flow through the inverse diodes as short as possible, and of keeping the drive current as small as possible when the MOSFETs are in the on-state so as to minimize power dissipation.